High-speed signaling systems and methods with adaptable, continuous-time equalization

ABSTRACT

A receiver includes a continuous-time equalizer, a decision-feedback equalizer (DFE), data and error sampling logic, and an adaptation engine. The receiver corrects for inter-symbol interference (ISI) associated with the most recent data symbol (first post cursor ISI) by establishing appropriate equalization settings for the continuous-time equalizer based upon a measure of the first-post-cursor ISI.

FIELD OF THE INVENTION

The present invention relates generally to the field of communications, and more particularly to high speed electronic signaling within and between integrated circuit devices.

BACKGROUND

The performance of many digital systems is limited by the interconnection bandwidth within and between integrated circuit devices (ICs). High performance communication channels between ICs suffer from many effects that degrade signals. Primary among them is inter-symbol interference (ISI) from high frequency signal attenuation and reflections due to impedance discontinuities.

ISI becomes more pronounced at higher signaling rates, ultimately degrading signal quality to the point at which distinctions between originally transmitted signal levels may be lost. Some receivers cancel ISI using a decision-feedback equalizer (DFE). DFEs multiply each of N recently received symbols by respective tap coefficients, the resulting products representing the ISI attributable to the corresponding symbol. The sum of these products is subtracted from the received signal prior to sampling. The ISI associated with the prior data is thereby reduced or eliminated.

In very high-speed systems it can be difficult to resolve the most recent data bit or bits in time to calculate their impact on the incoming symbol. Some receivers therefore ignore the impact of such symbols on the incoming signal, and consequently fail to correct for the ISI attributed to those symbols. Other receivers employ partial response DFEs (PrDFEs) that obtain multiple samples of the incoming data using multiple correction coefficients, one for each of the possible values of the most recently received symbol or symbols. The correct sample is then selected after the most recently received symbol or symbols are resolved.

PrDFEs are effective, but require a separate subtraction and sampling path for each possible value of the most recently received symbol or, in the case of multiple symbols (multi-symbol PrDFE), a separate computational path for each possible combination of the multiple symbol values. This results in e.g. 2^(M) paths in a binary PrDFE system that considers M prior symbols. The additional paths occupy area, require power, and slow signal rates by increasing the input capacitance of the receiver. There is therefore a need for power and area-efficient receivers capable of filtering incoming signals to cancel ISI from the most recently received symbol or symbols.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 depicts a receiver 100, in accordance with one embodiment, that receives information from a transmitter (not shown) via a high-speed communication channel 105.

FIG. 2 depicts adaptation engine 135 in accordance with one embodiment.

FIG. 3 details an embodiment of a tap-value generator 205 of FIG. 2 that generates a tap value using a sign-sign, least-mean-squared (LMS) algorithm.

FIGS. 4A-4D are waveform diagrams illustrating how tap-value generator 205 generates the values for taps α₀ (AGCadj) and α₁ (EQadj) in accordance with one embodiment.

FIG. 5 depicts three eye diagrams 500, 505, and 510 that illustrate the impact on an incoming signal Veq′ of adjusting signals AGCadj and EQadj.

FIG. 6 schematically depicts equalizer 120 of FIG. 1 in accordance with one embodiment.

FIG. 7 details an embodiment of variable capacitor 645 of FIG. 6.

FIG. 8 schematically depicts a bias-voltage generator 800 for use with equalizer 120 of FIG. 6.

DETAILED DESCRIPTION

FIG. 1 depicts a receiver 100, in accordance with one embodiment, that receives information from a transmitter (not shown) via a high-speed communication channel 105. In one embodiment, receiver 100 is instantiated on an integrated-circuit (IC) device and channel 105 provides differential signals RN and RP to a like-named differential input port of receiver 100 via a pair of pads 110. Channel 105 is AC coupled and includes a termination element 115 in this example. In other embodiments channel 105 is e.g. DC coupled, single ended, or optical. In embodiments adapted to communicate over optical channels, receiver 100 may include an integrated optical-to-electrical converter. Receiver 100 includes an analog, continuous-time equalizer 120, a decision-feedback equalizer (DFE) 125, data and error sampling logic 130, and an equalization-adaptation engine 135.

Equalizer 120 equalizes differential data signal RP/RN, conveyed from channel 105 to an input port of equalizer 120, to produce an equalized signal Veq on a like-named output port. (As with other designations herein, Veq refers both to a signal and a corresponding node or port; whether a given designation refers to a signal or a circuit element will be clear from the context.) Receiver 100 corrects for inter-symbol interference (ISI) associated with the most recent data symbol (first post cursor ISI) by establishing appropriate equalization settings for continuous-time equalizer 120 based upon a measure of the first-post-cursor ISI. In doing so, receiver 100 can eliminate the need to resolve the most recent data bit in time to calculate its impact on the incoming signal, and thus facilitate communication at higher speeds without the attendant complexity and power required by PrDFE-based receivers. Some embodiments may use PrDFE for subsequent filter taps or to complement the continuous-time equalizer.

Equalizer 120 amplifies signal RP/RN using a range of amplification factors, with higher frequency components typically being treated to higher amplification factors. Channel 105 will typically exhibit a low pass filter effect, in which case equalizer 120 may be used to compensate for attenuation of higher-frequency signal components. In some embodiments, the low-frequency gain of equalizer 120 may also be adjusted to compensate for broadband signal attenuation. Gain adjustments can be accomplished by frequency-selective amplification or attenuation, or a combination of amplification and attenuation. In general, the goal of equalization is to reduce or minimize the effects of ISI, so equalization is typically accomplished by adjusting one or more characteristics of a signal in a manner that mitigates the effects of ISI.

DFE 125 further equalizes signal Veq to produce a second equalized signal Veq′ for sampling logic 130. DFE 125 stores sequences of sampled data in a buffer 160 as post-tap data values. Though not shown, tap select logic may be included to enable selection of a subset of data values within buffer 160. Receive-side equalization taps can thus be selected to have latencies that match whatever ISI components are evident in channel 105. Each stored data value in buffer 160 after the initial latch is multiplied by a corresponding tap coefficient. The resulting products are summed and the total added to equalized signal Veq to produce the second equalized signal Veq′. In one embodiment clock signal DfeClk to DFE 125 is a recovered clock signal synchronized to the edges of the equalized signal as observed at the input of sampler 155. The DfeClk is phase offset from (e.g. the complement of) receive clock RClk. The error sampler can be timed to the edges of the equalized signal in other embodiments, as by tying the clock terminal of sampler 150 to an edge clock signal (not shown).

Amplifier 140 within sampling logic 130 compares signal Veq′ with a selected data level Dlev, outputting a signal indicative of a logic one (zero) if Veq′ is greater than (less than) level Dlev. Sampler 150 periodically captures the output from amplifier 140 on rising edges of a receive clock signal RClk to produce a series of error samples Err_(n). A second amplifier 145 compares signal Veq′ with a reference voltage Vr (e.g., zero volts), outputting a signal indicative of a logic one (zero) if Veq′ is greater than (less than) level Vr. Sampler 155 periodically captures the output from amplifier 145 on rising edges of receive clock signal RClk to produce a series of data samples Data_(n).

Adaptation engine 135 employs data and error samples Data_(n) and Err_(n) from sampling logic 130 to generate the tap values for equalizer 120 and DFE 125. In an embodiment in which equalizer 120 is adapted to provide both automatic gain control (AGC) to compensate for broadband gain and equalization to compensate for ISI, adaptation engine 135 generates measures of DC attenuation and one or more ISI values by comparing error signals Err_(n) with data samples of various symbol latencies. Based upon these generated values, adaptation engine 135 issues low-frequency control signals LFadj and high-frequency control signals HFadj to a control port of equalizer 120, and thereby controls the low-frequency gain and the peaking response of equalizer 120. In other embodiments a single control signal can control multiple equalization parameters, including e.g. the low-frequency gain and the peaking response,

Four simplified frequency-response diagrams 165, 170, 175, and 180 in the lower portion of FIG. 1 depict the approximate effects of adjusting the low-frequency and high-frequency gain of equalizer 120 in one embodiment. As shown in diagram 165, increasing the value of signal LFadj tends to increase the gain of equalizer 120 at low frequencies. With reference to diagram 170, increasing the value of signal HFadj tends to decrease the peak response of equalizer 120 around a particular (high) frequency of interest. Diagram 175 shows how the broadband frequency response of equalizer 120 is adjusted by moving signals LFadj and HFadj together in opposite directions. Diagram 180 shows how the equalization frequency response of equalizer 120 is adjusted by moving signals LFadj and HFadj together in the same direction. Equalizer 120 can equalize incoming signals by attenuating or amplifying some frequency components more than others, or by a combination of amplification and attenuation.

The LFadj signal from adaptation engine 135 adjusts the low-frequency gain of equalizer 120. The HFadj signal from adaptation engine 135, adjusts the peaking response of equalizer 120. Signals LFadj and HFadj are combinations of the α[1:0] signals that indicate the broadband gain (AGCadj) and equalization emphasis (EQadj) desired. The remaining adjustment signals α[N:2] are measures of the remaining ISI attributes due to the prior data symbols stored within buffer 160.

FIG. 2 depicts adaptation engine 135 in accordance with one embodiment. Adaptation engine 135 includes a series of synchronous storage elements 200 and tap-value generators 205 that together generate, from data and error samples Data_(n) and Err_(n), tap values α[1:0] for equalizer 120 and α[N:2] for DFE 125. The data and error samples are received on respective input ports, while the α values are conveyed to equalizer 120 and DFE 125 via the corresponding adaptation-engine output ports. Tap-value generators 205 each compare incoming error signals Err_(n) with either a current data sample Data_(n) or one of N−1 prior data samples to compute tap values α[N:0]. Element 210 shows the arithmetic logic utilized to generate LFadj and HFadj signals from AGCadj and EQadj (α[1:0]). Increasing the value of signal HFadj decreases the peaking response of equalizer 120 in this embodiment.

FIG. 3 details an embodiment of a tap-value generator 205 of FIG. 2 that generates a tap value using a sign-sign, least-mean-squared (LMS) algorithm. Generator 205 includes an XOR gate 300, logic 302 to convert the unsigned XOR output to a signed number, a multiplier 305 to scale the signed number by a constant μ, an adder 310, and a register 315. XOR gate 300 compares the corresponding data and error samples and presents its output to multiplier 305 via converter 302. The data and error samples represent the signs of the sampled values, so XOR gate 300 and converter 302 collectively have the effect of multiplying the signs and presenting the result to multiplier 305. Multiplier 305 multiplies the resulting product by a selected gain step size μ for the filter tap. Adder 310 adds the output from multiplier 305 to the current contents of register 315, which is then updated with the new count. Register 315 thus accumulates a count representative of the α value for the filter tap associated with the data samples of a particular latency. The α value for the filter tap is, in turn, representative of the ISI contribution of that filter tap to the present symbol. Ideally, each α value exactly offsets the respective ISI contribution. Perfection is difficult to obtain in practice, however, and the optimal tap values tend to vary with e.g. temperature and supply-voltage. Tap value generator 205 thus adaptively maintains representative α values that approximate the respective ISI contributions.

FIGS. 4A-4D are waveform diagrams illustrating how tap-value generator 205 generates the values for taps α₀ (AGCadj) and α₁ (EQadj) in accordance with one embodiment. Turning first to FIG. 4A, a signal trace 400 represents an incoming analog signal Veq′ over two symbol times t_(n−1) (the window for prior data Data_(n−1)) and t_(n) (the window for current data Data_(n)), in a case where signal conveys a data value of 1 at each symbol time. In this embodiment, Vr is equal to zero. Broadband gain adjustments are based upon the current sampled data value Data_(n) and the current sampled error value Err_(n). The sampled error is not shown; however, it can be seen that error sample Err_(n) for FIG. 4A would be zero because the value of trace 400 is less than Dlev in the time interval for t_(n). In that case, the AGCadj is incremented to increase the broadband gain of equalizer 120. The same holds true for the example of FIG. 4C. In FIGS. 4B and 4D, however, the current value of Veq′ is greater than Dlev, indicating that the sign of Err_(n) is one, in which case tap value AGCadj is decremented to reduce the broadband gain.

Returning to FIG. 4A, adjustments to EQadj are based upon the prior sampled data value D_(n−1) and the current sampled error value Err_(n). As noted previously, error sample Err_(n) for FIG. 4A is zero because the value of trace 400 is less than Dlev in the current time interval. Also evident in FIG. 4A, the value Veq′ for the prior sample time t_(n−1) is positive (i.e., D_(n−1)=1) because Veq′ is greater than reference voltage Vr (e.g., zero volts). In that case, the EQadj is incremented to simultaneously decrease the high-frequency and increase the low-frequency gain of equalizer 120. The high-frequency tap value EQadj is likewise incremented if the current error signal is a one and the prior data signal is a zero, as shown in FIG. 4D. On the other hand, EQadj is decremented, to simultaneously increase the high-frequency and decrease the low-frequency gain, if the current error sample has the same value as the prior data sample, conditions that are represented in FIGS. 4B and 4C.

The forgoing error comparisons are based upon the upper signal level defined by voltage Dlev and applied via amplifier 140. Adaptation engine 135 only updates the tap values α[N:0] based upon measurements that take place when the current data sample Data_(n) is a logic one. Adaptation engine 135 therefore includes a data filter, not shown, to prevent updates when the current sample Data_(n) is a logic zero. Other embodiments can include a second amplifier/sampler pair to generate error samples, such as by comparing the incoming signal Veq′ with the lower data level −Dlev, or the reference voltage to amplifier 140 can be varied over a number of values or ranges of values to facilitate additional testing and error-correction methods.

FIG. 5 depicts three eye diagrams 500, 505, and 510 that illustrate the impact on an incoming signal Veq′ of adjusting signals AGCadj and EQadj. Beginning with diagram 500, a signal eye 515 is of relatively low amplitude with respect to a desired data level Dlev. In this case, using the method described above in connection with FIGS. 4A-4D, the broadband gain of equalizer 120 may be increased to expand eye 515. With reference to diagram 505, the gain would continue to increase stepwise until eye 515 expanded such that signal level Dlev was in the center of the upper “fuzz” band 520. At the center of the fuzz band, the error sample (Err_(n)) from sampling logic 130 would exhibit an equal likelihood of sampling a one or a zero when the current data D_(n)=1, thus there would be no further net change in AGCadj.

We next consider the impact of adjusting value EQadj. Assuming DFE 125 is doing a reasonable job of cancelling the ISI associated with the post-cursor values for taps two through N, the remaining ISI at Veq′ contributing to the width of fuzz band 520 is assumed to be largely a result of first post-cursor ISI. Using the method described above in connection with FIGS. 4A-4D, the equalizer gain of equalizer 120 would be increased or decreased as necessary to reduce the amplitude of fuzz band 520. The adjustment would continue stepwise until eye fuzz band 520 diminished in the manner depicted in diagram 510 of FIG. 5. Thereafter the EQadj, the α1 tap, would experience an equal likelihood of incrementing and decrementing.

FIG. 6 schematically depicts equalizer 120 of FIG. 1 in accordance with one embodiment. Equalizer 120 includes two nearly identical stages 600 and 605, the second of which is depicted as a black box for ease of illustration. Other embodiments include more or fewer stages, or other circuit topologies with similar frequency responses. Equalizer stage 600 includes a pair of differential input transistors 615 and 620 with respective loads 625 and 630. Source degeneration is provided by a resistor 635, a transistor 640, and a pair of variable capacitors 645 and 650. The capacitance provided by transistors 645 and 650 is in parallel with resistor 635 and transistor 640 from a differential small-signal perspective, so the net impedance between the sources of transistors 615 and 620 decreases with frequency. As a consequence, the gain of equalizer stage 600 increases with frequency. The resistance through transistor 640 can be adjusted to change the source-degeneration resistance, and thus to alter the low-frequency response of stage 600. The capacitance through capacitors 645 and 650 can be selected to alter the peaking response (high frequency gain) of stage 600.

In an alternative embodiment, source degeneration is provided by one or more metal-insulator-metal (MIM) capacitors connected in parallel with resistor 635. The MIM capacitors can be used instead of or in addition to capacitors 645 and 650. Other control mechanisms might also be used to alter the source-degeneration resistance, as by digitally switching in different sizes and combinations of resistors. In still other embodiments the DC gain adjustment is supported via a separate gain-control amplifier, or is omitted altogether.

A DAC 655 converts the digital equalization setting LFadj[3:0] from e.g. adaptation engine 135 of FIG. 1 to a gate voltage for transistor 640. The value of the equalization setting thus determines the resistance between the sources of transistors 615 and 620, and consequently the low frequency gain of equalizer stage 600. In one embodiment, the output voltage from DAC 655 increases as setting LFadj [3:0] increases from 0000 to 1111. This maximum output represents the lowest resistance between the sources of transistors 615 and 620, and consequently the highest gain setting for stage 600. The output voltage of a similar DAC (not shown) in stage 605 performs a similar function as DAC 655 in stage 600.

FIG. 7 details an embodiment of variable capacitor 645 of FIG. 6: capacitor 650 is identical. Capacitor 645 includes a number of capacitor-connected transistors 700 and respective select transistors 705 controlled by signal HFadj. The areas, and thus the capacitances, of transistors 700 can vary from one to the next (e.g., their areas can be binary coded) for added granularity, or can be thermometer coded to reduce adjustment glitches that might otherwise occur when switching between values. Increasing values of HFadj[3:0] represent decreasing amounts of capacitance in the degeneration network, and therefore decreasing high-frequency gain.

FIG. 8 schematically depicts a bias-voltage generator 800 for use with equalizer 120 of FIG. 6. A resistor 805 and transistors 810 and 815 form a half-circuit replica of equalizer stage 600, with the input common-mode voltage Vin_com applied to the gate of transistor 810. A feedback loop including an amplifier 820 and a pair of transistors 825 and 830 sets the voltage on the inverting (−) terminal of amplifier 820 equal to the voltage applied to the non-inverting (+) terminal. In an embodiment in which supply voltage Vdd is 1.2 volts, a resistor divider provides one-volt to the non-inverting terminal of amplifier 820. The resulting bias voltage Vbias to stages 600 and 605 then establishes a one-volt common-mode output voltage for those stages. In some embodiments, lower common-mode voltages are avoided to ensure that transistors 615 and 620 of FIG. 6 are always in saturation.

In the foregoing description and in the accompanying drawings, specific terminology and drawing symbols are set forth to provide a thorough understanding of the present invention. In some instances, the terminology and symbols may imply specific details that are not required to practice the invention. For example, the interconnection between circuit elements or circuit blocks may be shown or described as multi-conductor or single conductor signal lines. Each of the multi-conductor signal lines may alternatively be single-conductor signal lines, and each of the single-conductor signal lines may alternatively be multi-conductor signal lines. Signals and signaling paths shown or described as being single-ended may also be differential, and vice-versa. Similarly, signals described or depicted as having active-high or active-low logic levels may have opposite logic levels in alternative embodiments.

A signal driving circuit is said to “output” a signal to a signal receiving circuit when the signal driving circuit asserts (or de-asserts, if explicitly stated or indicated by context) the signal on a signal line coupled between the signal driving and signal receiving circuits. The output (input) of a signal driving (receiving) circuit is generically referred to as an output (input) port. Circuit elements are controlled by application of control signals to respective control ports.

An output of a process for designing an integrated circuit, or a portion of an integrated circuit, comprising one or more of the circuits described herein may be a computer-readable medium such as, for example, a magnetic tape or an optical or magnetic disk. The computer-readable medium may be encoded with data structures or other information describing circuitry that may be physically instantiated as an integrated circuit or portion of an integrated circuit. Although various formats may be used for such encoding, these data structures are commonly written in Caltech Intermediate Format (CIF), Calma GDS II Stream Format (GDSII), or Electronic Design Interchange Format (EDIF). Those of skill in the art of integrated circuit design can develop such data structures from schematic diagrams of the type detailed above and the corresponding descriptions and encode the data structures on computer readable medium. Those of skill in the art of integrated circuit fabrication can use such encoded data to fabricate integrated circuits comprising one or more of the circuits described herein.

While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. For example, the depicted embodiments are signal-data-rate (SDR) systems, but other embodiments may support e.g. double-data-rate (DDR) or quad-data-rate (QDR) operation instead of or in addition to SDR operation. Furthermore, the receivers described above employ current-mode signaling, but might also be adapted to employ voltage-mode schemes in which signals are conveyed as modulated voltages. Voltage thresholds may also be employed in the latter case by simply converting current signals to voltage for comparison with a voltage reference. In addition, embodiments of the invention may be adapted for use with multi-pulse-amplitude-modulated (multi-PAM) signals, and PrDFE taps can be inserted after equalizer 120. Moreover, some components are shown directly connected to one another while others are shown connected via intermediate components. In each instance the method of interconnection, or “coupling,” establishes some desired electrical communication between two or more circuit nodes, terminals, or ports. Such coupling may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description. Where U.S. law applies, only those claims specifically reciting “means for” or “step for” should be construed in the manner required under the sixth paragraph of 35 U.S.C. § 112. 

What is claimed is:
 1. An integrated circuit comprising: a single-ended port to receive an input signal from a channel affected by intersymbol interference (ISI), including first-post-cursor ISI and subsequent post-cursor ISI; a continuous-time equalizer coupled to the single-ended port to produce a first equalized version of the input signal, the continuous-time equalizer including a first control port to control continuous-time equalization based upon a measure of the first-post-cursor ISI; and a decision-feedback equalizer coupled to the continuous-time equalizer, the decision-feedback equalizer to produce, from the first equalized version of the input signal, a second equalized version of the input signal, the decision-feedback equalizer including a second control port to control decision-feedback equalization based upon a measure of the subsequent post-cursor ISI.
 2. The integrated circuit of claim 1, further comprising: a first sampler to sample the second equalized version of the input signal to produce first samples; a second sampler to sample the second equalized version of the input signal to produce second samples; and an adaptation engine coupled to the first sampler and the second sampler to issue control signals to the first control port of the continuous-time equalizer responsive to the first samples and the second samples.
 3. The integrated circuit of claim 2, wherein the adaptation engine generates, from the first samples and the second samples, a tap value representative of first post-cursor inter-symbol interference in the input signal, and wherein the adaptation engine applies the tap value to the first control port.
 4. The integrated circuit of claim 3, wherein the adaptation engine generates, from the first samples and the second samples, a second tap value representative of second post-cursor inter-symbol interference for the input signal, and wherein the adaptation engine applies the second tap value to the second control port.
 5. The integrated circuit of claim 2, the continuous-time equalizer exhibiting a low-frequency gain and a peaking response, the adaptation engine to adjust the low-frequency gain responsive to comparisons between the first samples and the second samples.
 6. The integrated circuit of claim 2, the continuous-time equalizer exhibiting a low-frequency gain and a peaking response, the adaptation engine to adjust the low-frequency gain in dependence on the first samples and corresponding ones of the second samples.
 7. The integrated circuit of claim 1, the input signal comprising a series of symbols, the decision-feedback equalizer to reduce the intersymbol interference only from symbols other than the most-recently received symbol in the series of symbols.
 8. The integrated circuit of claim 1, wherein the continuous-time equalizer exhibits at least one of frequency-selective amplification and frequency-selective attenuation.
 9. A method comprising: receiving a single-ended input signal effected by intersymbol interference (ISI), including first-post-cursor ISI and subsequent post-cursor ISI; measuring the first-post-cursor ISI and the subsequent post-cursor ISI; applying continuous-time equalization to the single-ended input signal based on the measure of the first-post-cursor ISI to produce a first equalized continuous-time signal; applying decision-feedback equalization to the first equalized continuous-time signal based on the measure of the subsequent post-cursor ISI to produce a second equalized continuous-time signal.
 10. The method of claim 9, further comprising: sampling the second equalized continuous-time signal to produce first samples; and sampling the second equalized continuous-time signal to produce second samples; wherein measuring the first-post-cursor ISI includes applying a least-mean squared algorithm to the first samples and the second samples.
 11. The method of claim 10, further comprising measuring DC attenuation of the input signal using the least-mean squared algorithm.
 12. The method of claim 11, further comprising controlling the continuous-time equalization responsive to the measures of DC attenuation.
 13. The method of claim 11, further comprising generating, from the first samples and the second samples, a tap value representative of the first post-cursor ISI.
 14. The method of claim 13, further comprising generating, from the first samples and the second samples, a second tap value representative of second post-cursor ISI and applying the second tap value to the decision-feedback equalization.
 15. The method of claim 9, the input signal expressing a series of symbols, the continuous-time equalization reducing intersymbol interference associated only with a most-recently received symbol in the series of symbols.
 16. The method of claim 15, the decision-feedback equalization reducing the intersymbol interference only from symbols other than the most-recently received symbol in the series of symbols.
 17. The method of claim 9, the continuous-time equalization providing at least one of frequency-selective amplification and frequency-selective attenuation.
 18. An integrated circuit comprising: a single-ended channel to communicate an input signal effected by intersymbol interference (ISI), including first-post-cursor ISI and subsequent post-cursor ISI; a continuous-time equalizer coupled to the single-ended channel to produce a first equalized version of the input signal, the continuous-time equalizer including a first control port to control continuous-time equalization based upon a measure of the first-post-cursor ISI; a decision-feedback equalizer coupled to the continuous-time equalizer, the decision-feedback equalizer to produce from the first equalized version of the input signal a second equalized version of the input signal, the decision-feedback equalizer including a second control port to control decision-feedback equalization based upon a measure of the subsequent post-cursor ISI; and means for measuring the first post-cursor ISI and the subsequent post-cursor ISI.
 19. The integrated circuit of claim 18, wherein the means for measuring the first post-cursor ISI and the subsequent post-cursor ISI measures DC attenuation of the input signal. 